Issues Choosing Flyback Transformer Specs & General Direction Help

Question

I'm working on trying to figure out how to wind transformers myself and picking values etc. I currently have PQ26/25 Bobbins with PC95PQ26/25Z-12 ferrite cores (N95).

I have some smaller E cores somewhere as well but I'm not entirely sure what/where or their specs.

I'm trying to design a practical example of a 60W SMPS flyback power supply / 85-260VAC Input / 12.1VDC Output

Basically: 85-260VAC input 12.1VDC Output

I mostly followed along with https://techweb.rohm.com/knowledge/acdc/acdc_pwm/acdc_pwm01/1311 and came up with these values:

Vin 95V Vout 12.1V Vo 13.05V Vor 95V Duty Max 0.5 Ratio 7.28

Iout 5A Lmax 6A Ispk 24A Ls 4.18uH Lp 221.68uH Ippk 3.16A

The ferrite cores that I have say they have an AL-Value of 6520 ungapped and gapped 630 Which following the calculations I arrived at 6 turns for the primary. and a NI of 18.99

Most examples I've seen have turns from 30-50 on the primary side and then 6-20 on the secondary side. In order to get the Np turns to 30-40 and a similar NI value, that AL-value needs to be closer to 100.

Using a small amount of turns like that makes it impossible to get close to the ratio. I have no idea if this is alright or not. I feel like picking a different core with different material (N47) and an air gap might make the most sense - or machining down the center leg to get the AL-value I'm looking for.

Based on the graph from the spec sheet https://product.tdk.com/system/files/dam/doc/product/ferrite/ferrite/ferrite-core/catalog/ferrite_mz_sw_pq_en.pdf If I machine a center gap (on the center leg) to be about 2mm I should have an AL-value around 100 but I can't figure out how to calculate that. I'm also not sure how to figure out how big the gap is to achieve the 630AL-value (I can't find that particular product code anywhere and the specs)

One thing I did notice is that if I lower the Iout the number of required turns goes up (and so does the inductance for Ls and Lp)

I guess my questions are these:

1. Am I even remotely on the right track with my understanding (and/or) calculations?
2. Is there other important metrics I should be looking at?
3. Any direction or suggestions?

Edit:

After reading @Verbal Kint 's answer I went ahead and built the first part of the circuit (in my kitchen lab, since I have these things on hand), the AC to DC adapter. I omitted the EMI filter/inductor and used:

1. 600V 1μF safety cap
2. 400V 120μF cap
3. GBU806 Bridge Rectifier
4. 60W load (old incandescent light bulb)

I then hacked an old prototype board and plugged it into an isolation transformer.

Hooking up the leads to the positive side and negative side of the rectifier I seen this on the scope:

and this

This might not have been the best format for a question on here. It's easier when you can narrow it down to a specific item. What I really needed was general design direction. I think I got that and quite a bit more.

Answer 1

Designing a converter operated on the ac mains is quite a long task and requires several iterations to converge to something that you can actually build. I would first start by designing the front-end section, the one truly performing the conversion from ac to dc (always lowercase) as the downstream circuit remains a high-voltage dc-dc converter.

The front-end section will tell what maximum peak voltage the switching transistor will see in operation when you have 265 V rms (always insert a space) as an input voltage but, most importantly, what minimum or valley voltage the dc-dc will experience when the input is 85 V rms (VAC is not a unit). For instance, at low line, people usually accepts a 30-40% voltage ripple on the bulk capacitor (the bulky 400-V capacitor) meaning the real low voltage at which you must design the dc-dc converter is \$85\cdot\sqrt{2}\cdot(1-0.4)\approx 72\,\mathrm{V}\$ which is quite a low value. But, remember, the converter must deliver its nominal power at this low level otherwise you may see 100-Hz ripple on the output and perhaps the controller trips the overload protection in case the circuit can't deliver the power. This is something which is too often overlooked in commercial designs, especially considering the bulk capacitor capacitance reduction with age. You'll find all the formulas to design the bulk capacitor in my book or through this note freely available from my page. Once you have a value, run a quick simulation to check the voltage ripple is what is expected and, most importantly, what is the worst-case rms current in the capacitor as it will determine the type you need for a reliable operation:

Now that you have these important parameters on hand, you can start designing the dc-dc section, a 12-V/6-A converter. There many possible ways to design a flyback converter and I usually start backwards, with the \$BV_{DSS}\$ of the selected transistor affected by a derating factor. A derating factor is a safety margin that you take - e.g. 15% - and you make sure that in any circumstances, this margin is not violated: in case the optocoupler is broken, when there is a short circuit at the board terminals (and not at the output cable ends) etc. Usually, people apply a 20-15% derating factor (sometimes more) for MOSFETs and up to 100% for diodes. This last one can be reduced once the prototype is assembled and you test various faulty configurations in which - via snubbers for instance - you make sure that you keep at least 50%. See the below slide excerpted from my APEC 2011 seminar that you can download from my page, The Dark Side of Flyback Converter:

Once you are there, then you can start looking at the transformer basic specifications and, in particular, the turns ratio:

Here, the turns ratio is 1:085 with a primary inductance of 730 µH. Continuous conduction mode (CCM) will be ensured at low line while discontinuous conduction mode or DCM will be entered in high-line conditions. You can easily graph the point at which the transition theoretically occurs at nominal power:

The maximum peak current to design the transformer should be around 2.4 A considering a 10% margin. This value should be accepted at a 100-°C core, which is a worst-case. So your transformer should be designed for 730-µH primary inductance accepting up to 2.4 A at high temp. Without entering into design details, then you will have to select a core and turns numbers in the primary side to respect the following equation and stay away from saturation. This is for a single inductor or a flyback transformer: \$NB_{sat}A_e>L_pI_{p,max}\$ in which \$I_p\$ is the absolute maximum current the controller will authorize in a worst-case situation during which the transformer must not saturate.

A quick simulation using the flyback ready-made template free available from my webpage should tell us if an operating point is possible at the lowest input which is 72 V:

After a few seconds, the periodic operating point (POP) is found by Elements, the free demo version, delivers the operating waveforms and confirms the 12-V output loaded by a 2.4-ohm resistance:

From these plots, you can extract the rms primary- and secondary-side rms currents necessary to size your wires (\$F_{sw}\$ = 65 kHz for the simulation) but also add parasitics such as the leakage inductance and a real MOSFET model. But you may quickly reach the upper limit for the demo.